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Implementation of the measuring methods with commercial software applications should be developed in the development environment of that applications. If used instruments or equipments are not supported by the commercial applications, it should be developed. When commercial applications are used, it must be also paid for licensing. Mihajla Pupina b, Belgrade, Serbia, E-mail: obrad insimtel.

This paper will show the developed software and method for automated measuring antennas and antennas systems with integrated microwave receivers in wide frequency band. Automated method for measuring antenna radiation pattern The measured antenna is on the positioner which has capability of manual fine angle settings. The output of the antenna or integrated receiver is connected to the power meter which is connected to the computer with measuring software. Transmitting referent antenna is connected to the RF signal generator which is also connected to the computer with measuring software.

For each angle of the measured antenna, the measuring software sends commands to the signal generator to set desired signal frequencies. For each set frequency, received level from measured antenna is read and stored to the database. When the antenna radiation pattern with integrated receiver is measured at multiple frequencies in the case when the receiver output is in digital form, the measuring method which can be used is shown on the block diagram in Fig.. This measuring method is similar like previously described, except that output of the integrated receiver is in digital format [8][9].

In some cases receiving frequency channel should be set by measuring software computer. In both shown methods is used azimuth positioner with manual angle setting. The described methods can be used for Command and data files for Gnuplot are creating by the measuring software. Automated method for measuring radiation pattern of antenna with integrated receiver with output in digital format III. The Components are same and for other methods. Source code for instrument control is separated from software for measuring methods into library, which can be used for different measuring applications [1].

The basic request in design of this library was that the programming classes for instruments be grouped by type and function of the instrument, not by type of connection. The library was developed in C programming language for. In Fig. All instrument classes are inherited from abstract class Instrument which has methods Open and Close for opening and closing connection to instruments.

Those methods are implemented in inherited classes and depend from instrument type and connection type. All classes inherited from the class Instrument have attribute InstConnection which describe type and parameter for connection to the instrument. The components of the software for automated measuring The instruments control component is realized like class library where is for each type of instrument is created separately abstract class.

The software for some measuring method can use any available instrument which is supported by library class and software developing can be focused on measuring method not on controlling and connecting with the instruments. The results of measuring are stored in database for later processing and presentation. It is usually for each measuring method to use two database tables. The first is used for storage data about measuring conditions, and the second to storage measured data. The measured data from database can be shown in graphical form or exported to Excel or other file format for later analysis or using.

In this stage of software development, it is used external software Fig. The abstract class Instrument and inherited classes for instrument types The classes for particular instrument type by function have methods which are specified for that instrument type. Implementations of those methods are in inherited classes for each instrument model. For each models the structure SigGenSpec has characteristics of the particular instrument model e. When measuring software is developed it will be made reference to base class for instrument type, not for particular instrument model.

It is also possible to list available instrument models, and with static method of the base class it is possible to create object of the selected instrument model. In this example with signal generator, static method CreateSigGen is defined for this purpose.

In this way is possible to change DLL instrument library with new one which support new instrument model without changing and compiling existing measuring software. For simplicity, in this diagram are not shown transitions to error states. The abstract class SignalGenerator and inherited classes for different instrument models Not all instrument models should have connection to the real equipment. For example, the class SigGenDummy is virtual instrument and it is very useful during measuring software development. Complete software was written in programming language C.

Measuring software application supports measuring for different methods. For each measuring methods were developed different customized windows forms. The software makes possible setup parameters for each measuring and choice of measuring instrument models. Configuration data are stored in XML files. Forms for application configuration are same for all measuring methods.

Windows form for selection and confguration instruments In the shown software, it is used local. Measured data in the antenna radiation pattern measuring method are stored into two database tables MeasAngle and MeasAngleResults. In MeasAngle table are saved general data about measuring like frequency range and step, time of start measuring, comment, etc. Software module for executing antenna radiation pattern measuring with integrated receivers with digital outputs is realized to use in advanced defined states.

Diagram states of the software modul for antenna radiation pattern measuring with integrated receivers wth digital outputs When the software module is started initialization of the communications with instruments and receiver and opening database will be done.

Signal generator frequency and receiver channel are incremented while boundary frequency is not reached. On Fig. The software helps to make measurement for shorten time and to present on convenient way large amount of measured data. Different presentation types of he measured data can help to check for possible problems in realization of the tested antennas. The software also can be upgraded for supports for a new instruments types and models, for different kinds of integrated receivers and to support new measuring methods.

Windows forms for antenna radiation pattern measuring VI. Because there are large number of measured data, tabular view is not always suitable. The application supports export measured data to Excel and CSV file formats for analysis or representing in other programs. Graphical views of the measured results are more convenient. Orthogonal, polar and 3D graphical views are supported for representing antenna radiation patterns.

On this diagram is easy to observe some irregularities in antenna radiation patterns on some frequencies. Kruft, P. Ponce-Cruz, F. Pirola, V. Teppati, V. Williams, C. Kelley, Gnuplot 4. Albahari, B. Albahari, C 3. Example of 3D antenna radiation pattern digram which depends from frequency Dipole antenna, often used in experimental setup to measure the level of electromagnetic field at some characteristic points in enclosure, is described by using a compact wire model implemented into transmission-line matrix TLM method.

Using the proposed model, impact of receiving antenna on shielding effectiveness is illustrated on several examples of enclosure with apertures and compared with the corresponding numerical and circuital approaches in which the antenna presence is neglected. Keywords Enclosure, shielding effectiveness, dipole antenna, wire TLM model.

The shielding enclosure performances are quantified by shielding effectiveness SE , defined as the ratio of field strength in the presence and absence of the enclosure. Integral parts of the shielding enclosure are apertures of various forms, intended for heat dissipation, control panels, outgoing or incoming cable penetration, airing or other purposes. The shielding enclosure with apertures should be designed based on the analysis of the EM coupling mechanism through apertures, in order to minimize the EM interference EMI and susceptibility risk due to inevitable discontinuities.

There are many techniques used to calculate SE, from analytical methods to numerical simulations. Analytical formulations [1] are a quick tool based on the Fourier transformation and the model analogy. Circuital approach has been modified in [4] to allow for considering oblique incidence and polarization and not to be limited by the location of aperture with respect of plane wave propagation direction.

In that model, the SE of the enclosure with apertures on multiple sides can be simply calculated by superposition of one dimensional result, and the problem of SE considering the field with arbitrary incidence and polarization angle can be solved by vector decomposition. Differential numerical techniques in the time domain, such as the Finite-Difference Time-Domain FDTD method [5] and Transmission Line Matrix TLM method [6], owing to their characteristics, have found their application in solving many electromagnetic compatibility EMC problems in a wide frequency range.

In [7] TLM method has been successfully employed to calculate SE of shielding enclosure with apertures over a broad frequency band up to 3 GHz. In parallel with this research, authors of this paper have conducted their own analysis of influence of various factors, such as aperture patterns, their dimensions, number and orientation with the respect of enclosure walls or plane wave propagation direction, on shielding efficiency of enclosure and the results have been presented in [8] and [9].

In addition, impact of plane wave excitation parameters of shielding properties of enclosure with multiple apertures has been considered by the authors in [10] and [11]. Again, TLM method was used in [8]-[11] to numerically study these various effects over a frequency range up to GHz. In practice, when EMC measurements are performed, to experimentally characterize the SE of enclosure, small dipole receiving antenna, is located inside the enclosure.

Such antenna is used to measure the level of EM field, coming from external interference source through apertures, at the points in the enclosure in order to perform the SE calculation. Receiving antenna of finite dimensions could significantly affect the EM field distribution inside the enclosure [1] and thus affect the results for SE.

Both either circuital or numerical approaches mentioned above did not take into account the presence of receiving antenna. Therefore, in this paper TLM method enhanced with compact wire model [13] to efficiently describe the dipole antenna, is applied in order to create a numerical model that can used to investigate the impact of receiving antenna on SE of enclosure. This model has been used here to calculate the SE of rectangular enclosure with one or two apertures of rectangular crosssection on the front wall and two adjacent walls, in the frequency range of up to GHz.

Obtained numerical results illustrate the SE variation due to receiving antenna in comparison with the circuital or numerical case when its presence is neglected. The simplest solution is to model wires by using short-circuit nodes or shorted link-lines adjacent to the wire surface [6]. This, however, is rarely a practical proposition, as computational resource limitation and geometrical disparity between the modelled space and fine features in the EMC problems result in a rather crude rectangular shape model of the wire.

Sophisticated solution in the form of compact wire model or wire node, which can allow for accurate modelling of wires with a considerably smaller diameter than the node size, has been introduced in TLM method [13]. It use special wire network embedded within TLM nodes Fig. In order to achieve consistency with the rest of the TLM model, the each segment of wire network within one TLM node is formed by using additional link and stub lines Fig.

The parameters of link and stub lines are chosen to model the per-unit length wire capacitance and inductance, while at the same time maintaining synchronism with the rest of the transmission line network. For an example, for the node containing i-directed straight wire segment, as depicted in Fig. Empirically found factors k Ci and k Li for the wire located in free space are: k Ci 0. Wire per-unit length capacitance is then modelled by the link line of characteristic impedance Z wi : t Z wi ' i Cwi 10 while the wire per-unit length inductance is modelled by shortcircuit stub of characteristic impedance Z wsi : i Z wsi L wi Z wi t 11 Fig.

The front wall of the enclosure was made of the 3 mm conductive material with rectangular apertures having different dimensions and numbers: one 50 mm x 10 mm aperture, one 50 mm x 30 mm aperture and two 50 mm x 10 mm apertures. We have calculated the enclosure SE for a plane wave of normal incidence to the front wall and vertical z electric polarization as excitation, as shown in Fig. Link and stub lines network for straight wire segment running in i direction The single column of TLM nodes, through which wire conductor segments pass, can be used to approximately form the fictitious cylinder which represents capacitance and inductance of wire per unit length.

Its effective diameter, different for capacitance and inductance, can be expressed as a product of factors empirically obtained by using known characteristics of TLM network and the mean dimensions of Fig. Rectangular enclosure with a rectangular aperture In the first case, for calculating SE using the numerical model, we assumed that the presence of the receiving antenna could be neglected empty enclosure. The SE was calculated at the point of the enclosure mm, mm, 60 mm for all specified patterns of rectangular apertures, by using the conventional TLM method Fig.

Next, we presented numerical results obtained when the receiving antenna was not included in the model, when the receiving antenna was included in the model, and the modified circuital model results described in [4]. The calculated SE of the receiving dipole antenna at the centre point mm, mm and 60 mm and of the empty enclosure at the same point, are shown in Figs. Numerical results for the SE of enclosure without antenna at the point mm x mm x 60 mm As can be seen from Fig. This indicates that the patterns and the number of apertures only affect the level of attenuation to which EM field propagating through apertures is exposed.

As expected, the level of SE decreases with the increase of area covered by apertures. However, as it will be shown next, these results deviate to some extent from the case when the receiving antenna is included in the model. For calculating SE when the receiving dipole antenna is included in the numerical model of the enclosure by using the compact wire model described in section II, the antenna is modelled as z-directed 80 mm long wire having the diameter of 1. Its position within the enclosure is defined by points mm, mm, 0 mm and mm, mm, mm.

The numerical results for the SE of enclosure, shown in Fig. In comparison with the case when enclosure is empty, it can be seen that the antenna presence significantly decreases the SE of enclosure in the whole frequency range. Results for SE of enclosure, with one aperture of dimension 50 mm x 10 mm, at the point mm x mm x 60 mm Fig. Results for SE of enclosure, with one aperture of dimension 50 mm x 30 mm, at the point mm x mm x 60 mm Fig. Numerical results for the SE of enclosure with antenna at the point mm x mm x 60 mm Fig. Results for SE of enclosure, with two apertures of dimension 50 mm x 10 mm, at the point mm x mm x 60 mm It can be seen that the presence of the receiving antenna significantly reduces the shielding efficiency of the enclosure, as the SE level is always lower in comparison with the case when the enclosure is empty.

Although the presence of antenna is neglected in the circuital approach, the SE values of enclosures with one and two 50 mm x 10 mm apertures are lower than the numerical results obtained. For the enclosure with one 50 mm x 30 mm aperture dimensions, the results obtained by circuital approach practically coincide with the numerical results obtained for an empty enclosure.

Also, there is a tendency to slightly shift the resonant frequencies of enclosure. Finally, the rectangular enclosure with apertures on multiple sides for oblique incident plane wave is considered. Numerical results for SE at the point mm x mm x 60 mm obtained by using TLM method with and without compact wire model and results obtained using circuital approach at the same point are shown in Fig. Two groups of two rectangular apertures of dimension 50 mm x 10 mm are placed on the adjacent enclosure walls while an obliquely incident wave with the azimuth angle 60, elevation angle 90, and polarization angle 30 is used as excitation.

It can be seen that the presence of the receiving antenna significantly reduces the SE of the enclosure and that results obtained using circuital approach are lower than the numerical results. Results for SE of enclosure, with two groups of two rectangular apertures of dimension 50 mm x 10 mm placed on the adjacent enclosure walls, at the point mm x mm x 60 mm IV. The given example confirm that the antenna presence affects the EM field distribution inside the enclosure and thus affects the SE level results, as well as the location of resonant frequencies.

The results of circuital approach do not coincide in all cases with the numerical simulation results, but this fast analytical method can be used for approximate calculation of SE of enclosures. Future research will comprise numerical estimation of antenna impact on SE for given antenna dimensions and also modification of the circuital model in order to include the presence of the receiving antenna, since it is a real element in the measurement process. Park and H. Eom, Electromagnetic penetration into a rectangular cavity with multiple rectangular apertures in a conducting plane IEEE Trans.

EMC-0, no. Robinson, T. Benson, C. Christopoulos, J. Dawson, M. Ganley, A. Marvin, S. Porter, D. Shim, D. Kam, J. Kwon, J. Kunz, R. Nie, P. Du, Y. Yu, Z. Scientific- Professional Symp. B-I-6, pp. Milutinovic, T. Cvetkovic, N. Doncov, B. Milovanovic, Analysis of enclosure shielding properties dependence on aperture spacing and excitation parameters, in Proc. IEEE Conf. Wlodarczyk, V. Trenkic, R. Scaramuzza, C. The aim of this paper is to analyze transmission parameters such as propagation loss, diffraction fading, attenuation due to atmospheric conditions etc.

Reliability of radio-relay links, depending on bandwidth channel, digital modulation types and bit rate, is considered using the simulation programme Pathloss. Annual rain and multipath unavailability are analyzed at the frequency of 18 GHz band on two links and four bandwidth channels. These results are particularly important in the design of transmission links over Wireless Internet. Keywords Radio-relay, reliability, rain unavailability, multipath I. These methods have been developed for the purpose of predicting the most important propagation parameters for radio-relay links [1].

In design of line-of-sights of radio-relay links several propagation parameters must be taken into consideration. These parameters are: diffraction fading due to an obstacle on terrain path, attenuation in the atmosphere, fading due to atmospheric multipath, fading due to multipath arising from the surface reflection, attenuation due to precipitation, variation of the angle-of-launch at the transmitter terminal and the angle-of-arrival at receiver terminal due to refraction, reduction in cross-polarization discrimination in multipath or precipitation conditions and signal distortion due to frequency selective fading and delay during multipath propagation.

The probability of appearance of these events is of great importance in consideration of reliability of line-of-sight radio-relay systems. All these parameters are calculated in designing of terrestrial radio-relay links to points which are used to provide wireless Internet.

From these points wireless Internet is distributed at standardized frequencies to end users. The latter is important for obtaining high even harmonic rejection ratios. When LO f LO max f no frequency division is used and we rely solely on the symmetry of the VCOs, therefore even harmonic rejection is poor. However, at the highest receiving frequencies the harmonic mixing presents less concern. The delay needed for the 5th order HR is implemented by controlled delay cells and an appropriate design of the frequency divider.

The corresponding delay cells are connected to appropriate outputs of the divider depending on the VCO division ratio used.

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When f LO f LO max 10, the needed delay value is obtained exactly without using delay cells. The control voltage for the delay cells is produced in a DLL with a delay line consisting of 10 delay cells of the same type as DCa and DCb. CLK duty cycle error, timing errors due to shift registers and delay cell, as well as amplitude mismatches are assumed to be uncorrelated, normally distributed random variables with standard deviations, DtC, ShR and DU respectively.

A We assumed that ShR 0. After summing up the two equations above, we obtain kq pl cos P. Therefore cos P Q. We found, that cos PQ. But cos P sin P. Obviously P 1 PQ. Therefore cos P can take values 0, 1 and 1. As a result, the minimal non-zero value of P is 3. Hence P 6 must be fulfilled contradicting our requirements. As it was expected, the HRRs worsen significantly for the highest LO frequencies when no frequency divider is used. In addition, relative time errors decrease when the LO period rises. As a result, the HRR s are significantly increased.

LO Additionally, no weighting is used. The penalty is some HR degradation at highest LO frequencies. On the other hand HRR can be significantly improved if a digital cancellation technique is applied, similar to those in [3]. Do, C. Efficient Estimation of the Antenna Noise Level Using Neural Networks Ivan Milovanovic 1, Zoran Stankovic and Marija Milijic 3 Abstract This paper presents how through the use of artificial neural networks we can accelerate the prediction procedure of the external noise level at the receiving point of wireless communication systems.

Were taken into account only the effects of natural noise sources, which are surrounded by the antenna system and considerably more stable than artificial. The case of microwave wireless transmission, where dominated influence of noise generated by emissions of gases from the atmosphere primarily oxygen and water vapour , is considered. Accordingly, we developed a neural network model for antenna noise temperature prediction of the RF receiver based on Multilayer Perceptron MLP network.

The architecture of this model, the results of its training and testing and simulation results are presented in this paper in the appropriate sections. Keywords Neural network, Antenna Noise, Brightness temperature. The wireless system design goal is to achieve the largest possible coverage area in which the received power is sufficiently strong compared to background noise. Consequently, one of fundamental parameters in wireless communication is signalto-noise power ratio that indicates the reliability of the link between the transmitter and receiver.

Therefore, it certainly helps to have a reliable tool to estimate noise power during the process of wireless systems designing. Many noise dependences [1] are represented by formula whose parameters should be determined from a lot of complex figures. ANN is very sophisticated modeling techniques capable of modeling extremely complex functions. Indeed, anywhere that there are problems of prediction, classification or control, neural networks can be introduced. ANN has the capability of a functional dependence s modeling exclusively on the basis of input data [-5].

Neural network architecture which is consisted of connected small processing units neurons. In this way, neural network can be used for modeling high-distributed and high-parallel problems [-5]. The second is neural network ability to learn function dependence on the basis of solved examples rather then to learn to execute some well known function dependence.

After successful learning process of neural network, it can be used not only for known examples but also for unknown examples generalization. Neural network has been used for estimation level of RF receiver external noise versus only frequency, without taking into account the parameters that describe the antenna environment [4]. In this paper, neural model for prediction temperature of noise source brightness is developed resulting in more effective estimation of receiver external noise dependence on frequency and antenna elevation in microwave range.

Both internal and external noise must be considered. The available noise power is obtained by summing the contributions of each individual noise sources. To be able to perform the calculation it is necessary to introduce a parametar that determines the noise radiation sources. The parameter used in that sense commonly is brightness [6,7]. Cosmic noise decreases approximately with the square of the frequency so that the above 1 GHz is very small and can be ignored by receiver operating in the microwave range.

Noise from Earth, that correlates average noise temperature of 54 K, is important only for satellite antenna with the main radiation bean directed to Earth. There are number noises from many cosmic objects, but the only significant is the noise from the Sun. The Sun noise significantly affects on antenna noise only when large direction antenna with main radiation beam directed to the Sun. Atmospheric noise can derive from two sources. In The first is electrostatic discharge in atmosphere that overcomes for frequency range bellow 50 MHz. The last is emission in atmosphere due to water vapor and oxygen that is dominant in high frequency range Figure 1.

Temperature of atmosphere brightness versus antenna elevation and frequency when average concentrate of tropopause water vapor is 7. For large space angle of antenna radiation, antenna noise temperature is approximately equal as temperature of antenna brightness from atmosphere that radiates noise.

Also, it considers calm and good weather with constant atmosphere condition with average concentrate of water vapor is 7. For given conditions, brightness temperature depends on antenna elevation angle and frequency. If weight matrice w is presented as matrix structure, it can cause difficulties in implementation neural network and in its training algorithm. For this reason, neural network weight matrice w is replaced by set of neural network weights whose elements are weight matrices and vector of biases of neural network layers. During process of 9. The figure.

The vector of l-th hidden layer outputs can be presented using vector y l with dimension N l 1 where N l is number of neurons in l-th layer. The training of neural model is done using samples that are visual read from the graphics Figure 1. The samples are read in frequency range 1. GHz f Levenberg-Marquartd method is used for training neural model with accuracy The architecture of MLP neural model of antenna brightness temperature versus atmosphere in microwave range while the atmosphere conditions are constant All neurons from the last hidden layer H are connected with the neuron of the output layer.

Scattering diagram for MLP model The test of every trained MLP model is done with the set of 48 samples that are read in frequency 1. This dependence is got for less then 4secunds using Pentium IV 1. Classic way of visual reading from different ITU recommendation can be time consuming and with great error possibility because of visual reading and applying interpolation formulas.

The good alternative can be neural networks model of very complex graphs from ITU recommendation. Neural network model can avoid errors due to manual graphs reading enabling faster calculation of the level of external noise of receiver Neural model also enables the automation of the process of predicting noise power of receiver making one suitable method for the efficient analysis of the entire coverage area of wireless communication system transmitters in a big number of points that is of vital importance for the design and analysis of all components of modern wireless communication systems.

Figure 3. It can be seen very satisfying agreement between neural model output and samples that are visual read from the graphics in figure 1.

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The model MLP is used for simulation of antenna brightness temperature caused by atmosphere that radiates noise versus antenna elevation and frequency. It can be seen very satisfying agreement between these results and referent values proving the choice of this model. Figure 5 presents 3D dependence of antenna brightness temperature versus atmosphere that radiates noise versus antenna elevation and frequency using points per frequency x 91 per [1] Recommendation ITU-R P.

Hertz, A. Krogh, R. Jun- 1. Jul , pp. Zentner, Antene i radiosustavi, Graphis, Zagreb This software can be used with all antennas types and antennas with integrated microwave receivers with analog or digital output. It is also shown results of measuring the horn antennas system with the integrated microwave receivers in frequency range from 15 to 19 GHz.

This capability helps to create very complex measuring methods which significantly accelerate and simplify measuring, especially if are used a lot of calibration data [1]. It is usual to develop or adopt software for each measuring methods []. There are many commercial software applications that can be used in the measuring automation like LabView from National Instruments [3][4], Matlab from Mathworks [5] or OpenLab from Agilent [6]. Implementation of the measuring methods with commercial software applications should be developed in the development environment of that applications.

If used instruments or equipments are not supported by the commercial applications, it should be developed. When commercial applications are used, it must be also paid for licensing. Mihajla Pupina b, Belgrade, Serbia, E-mail: obrad insimtel. This paper will show the developed software and method for automated measuring antennas and antennas systems with integrated microwave receivers in wide frequency band.

Automated method for measuring antenna radiation pattern The measured antenna is on the positioner which has capability of manual fine angle settings. The output of the antenna or integrated receiver is connected to the power meter which is connected to the computer with measuring software. Transmitting referent antenna is connected to the RF signal generator which is also connected to the computer with measuring software. For each angle of the measured antenna, the measuring software sends commands to the signal generator to set desired signal frequencies. For each set frequency, received level from measured antenna is read and stored to the database.

When the antenna radiation pattern with integrated receiver is measured at multiple frequencies in the case when the receiver output is in digital form, the measuring method which can be used is shown on the block diagram in Fig.. This measuring method is similar like previously described, except that output of the integrated receiver is in digital format [8][9].

In some cases receiving frequency channel should be set by measuring software computer. In both shown methods is used azimuth positioner with manual angle setting. The described methods can be used for Command and data files for Gnuplot are creating by the measuring software. Automated method for measuring radiation pattern of antenna with integrated receiver with output in digital format III.

The Components are same and for other methods. Source code for instrument control is separated from software for measuring methods into library, which can be used for different measuring applications [1]. The basic request in design of this library was that the programming classes for instruments be grouped by type and function of the instrument, not by type of connection. The library was developed in C programming language for.

In Fig. All instrument classes are inherited from abstract class Instrument which has methods Open and Close for opening and closing connection to instruments. Those methods are implemented in inherited classes and depend from instrument type and connection type. All classes inherited from the class Instrument have attribute InstConnection which describe type and parameter for connection to the instrument.

The components of the software for automated measuring The instruments control component is realized like class library where is for each type of instrument is created separately abstract class. The software for some measuring method can use any available instrument which is supported by library class and software developing can be focused on measuring method not on controlling and connecting with the instruments.

The results of measuring are stored in database for later processing and presentation. It is usually for each measuring method to use two database tables. The first is used for storage data about measuring conditions, and the second to storage measured data. The presentation component is used to show raw or processed measuring data in desired form. The measured data from database can be shown in graphical form or exported to Excel or other file format for later analysis or using. In this stage of software development, it is used external software Fig. The abstract class Instrument and inherited classes for instrument types The classes for particular instrument type by function have methods which are specified for that instrument type.

Implementations of those methods are in inherited classes for each instrument model. For each models the structure SigGenSpec has characteristics of the particular instrument model e. When measuring software is developed it will be made reference to base class for instrument type, not for particular instrument model. It is also possible to list available instrument models, and with static method of the base class it is possible to create object of the selected instrument model.

In this example with signal generator, static method CreateSigGen is defined for this purpose. In this way is possible to change DLL instrument library with new one which support new instrument model without changing and compiling existing measuring software. For simplicity, in this diagram are not shown transitions to error states. The abstract class SignalGenerator and inherited classes for different instrument models Not all instrument models should have connection to the real equipment.

For example, the class SigGenDummy is virtual instrument and it is very useful during measuring software development. Complete software was written in programming language C. Measuring software application supports measuring for different methods. For each measuring methods were developed different customized windows forms. The software makes possible setup parameters for each measuring and choice of measuring instrument models.

Configuration data are stored in XML files. Forms for application configuration are same for all measuring methods. Windows form for selection and confguration instruments In the shown software, it is used local. Measured data in the antenna radiation pattern measuring method are stored into two database tables MeasAngle and MeasAngleResults.

In MeasAngle table are saved general data about measuring like frequency range and step, time of start measuring, comment, etc. Software module for executing antenna radiation pattern measuring with integrated receivers with digital outputs is realized to use in advanced defined states. Diagram states of the software modul for antenna radiation pattern measuring with integrated receivers wth digital outputs When the software module is started initialization of the communications with instruments and receiver and opening database will be done.

Signal generator frequency and receiver channel are incremented while boundary frequency is not reached. On Fig. The software helps to make measurement for shorten time and to present on convenient way large amount of measured data. Different presentation types of he measured data can help to check for possible problems in realization of the tested antennas. The software also can be upgraded for supports for a new instruments types and models, for different kinds of integrated receivers and to support new measuring methods. Windows forms for antenna radiation pattern measuring VI.

Because there are large number of measured data, tabular view is not always suitable. The application supports export measured data to Excel and CSV file formats for analysis or representing in other programs. Graphical views of the measured results are more convenient. Orthogonal, polar and 3D graphical views are supported for representing antenna radiation patterns.

On this diagram is easy to observe some irregularities in antenna radiation patterns on some frequencies. Kruft, P. Ponce-Cruz, F. Pirola, V. Teppati, V. Williams, C. Kelley, Gnuplot 4. Albahari, B. Albahari, C 3. Example of 3D antenna radiation pattern digram which depends from frequency Dipole antenna, often used in experimental setup to measure the level of electromagnetic field at some characteristic points in enclosure, is described by using a compact wire model implemented into transmission-line matrix TLM method.

Using the proposed model, impact of receiving antenna on shielding effectiveness is illustrated on several examples of enclosure with apertures and compared with the corresponding numerical and circuital approaches in which the antenna presence is neglected. Keywords Enclosure, shielding effectiveness, dipole antenna, wire TLM model. The shielding enclosure performances are quantified by shielding effectiveness SE , defined as the ratio of field strength in the presence and absence of the enclosure.

Integral parts of the shielding enclosure are apertures of various forms, intended for heat dissipation, control panels, outgoing or incoming cable penetration, airing or other purposes. The shielding enclosure with apertures should be designed based on the analysis of the EM coupling mechanism through apertures, in order to minimize the EM interference EMI and susceptibility risk due to inevitable discontinuities.

There are many techniques used to calculate SE, from analytical methods to numerical simulations. Analytical formulations [1] are a quick tool based on the Fourier transformation and the model analogy. Circuital approach has been modified in [4] to allow for considering oblique incidence and polarization and not to be limited by the location of aperture with respect of plane wave propagation direction. In that model, the SE of the enclosure with apertures on multiple sides can be simply calculated by superposition of one dimensional result, and the problem of SE considering the field with arbitrary incidence and polarization angle can be solved by vector decomposition.

Differential numerical techniques in the time domain, such as the Finite-Difference Time-Domain FDTD method [5] and Transmission Line Matrix TLM method [6], owing to their characteristics, have found their application in solving many electromagnetic compatibility EMC problems in a wide frequency range. In [7] TLM method has been successfully employed to calculate SE of shielding enclosure with apertures over a broad frequency band up to 3 GHz. In parallel with this research, authors of this paper have conducted their own analysis of influence of various factors, such as aperture patterns, their dimensions, number and orientation with the respect of enclosure walls or plane wave propagation direction, on shielding efficiency of enclosure and the results have been presented in [8] and [9].

In addition, impact of plane wave excitation parameters of shielding properties of enclosure with multiple apertures has been considered by the authors in [10] and [11]. Again, TLM method was used in [8]-[11] to numerically study these various effects over a frequency range up to GHz.

In practice, when EMC measurements are performed, to experimentally characterize the SE of enclosure, small dipole receiving antenna, is located inside the enclosure. Such antenna is used to measure the level of EM field, coming from external interference source through apertures, at the points in the enclosure in order to perform the SE calculation.


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Receiving antenna of finite dimensions could significantly affect the EM field distribution inside the enclosure [1] and thus affect the results for SE. Both either circuital or numerical approaches mentioned above did not take into account the presence of receiving antenna. Therefore, in this paper TLM method enhanced with compact wire model [13] to efficiently describe the dipole antenna, is applied in order to create a numerical model that can used to investigate the impact of receiving antenna on SE of enclosure.

This model has been used here to calculate the SE of rectangular enclosure with one or two apertures of rectangular crosssection on the front wall and two adjacent walls, in the frequency range of up to GHz. Obtained numerical results illustrate the SE variation due to receiving antenna in comparison with the circuital or numerical case when its presence is neglected.

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The simplest solution is to model wires by using short-circuit nodes or shorted link-lines adjacent to the wire surface [6]. This, however, is rarely a practical proposition, as computational resource limitation and geometrical disparity between the modelled space and fine features in the EMC problems result in a rather crude rectangular shape model of the wire. Sophisticated solution in the form of compact wire model or wire node, which can allow for accurate modelling of wires with a considerably smaller diameter than the node size, has been introduced in TLM method [13].

It use special wire network embedded within TLM nodes Fig. In order to achieve consistency with the rest of the TLM model, the each segment of wire network within one TLM node is formed by using additional link and stub lines Fig. The parameters of link and stub lines are chosen to model the per-unit length wire capacitance and inductance, while at the same time maintaining synchronism with the rest of the transmission line network. For an example, for the node containing i-directed straight wire segment, as depicted in Fig. Empirically found factors k Ci and k Li for the wire located in free space are: k Ci 0.

Wire per-unit length capacitance is then modelled by the link line of characteristic impedance Z wi : t Z wi ' i Cwi 10 while the wire per-unit length inductance is modelled by shortcircuit stub of characteristic impedance Z wsi : i Z wsi L wi Z wi t 11 Fig. The front wall of the enclosure was made of the 3 mm conductive material with rectangular apertures having different dimensions and numbers: one 50 mm x 10 mm aperture, one 50 mm x 30 mm aperture and two 50 mm x 10 mm apertures.

We have calculated the enclosure SE for a plane wave of normal incidence to the front wall and vertical z electric polarization as excitation, as shown in Fig. Link and stub lines network for straight wire segment running in i direction The single column of TLM nodes, through which wire conductor segments pass, can be used to approximately form the fictitious cylinder which represents capacitance and inductance of wire per unit length.

Its effective diameter, different for capacitance and inductance, can be expressed as a product of factors empirically obtained by using known characteristics of TLM network and the mean dimensions of Fig. Rectangular enclosure with a rectangular aperture In the first case, for calculating SE using the numerical model, we assumed that the presence of the receiving antenna could be neglected empty enclosure.

The SE was calculated at the point of the enclosure mm, mm, 60 mm for all specified patterns of rectangular apertures, by using the conventional TLM method Fig. Next, we presented numerical results obtained when the receiving antenna was not included in the model, when the receiving antenna was included in the model, and the modified circuital model results described in [4].

The calculated SE of the receiving dipole antenna at the centre point mm, mm and 60 mm and of the empty enclosure at the same point, are shown in Figs. Numerical results for the SE of enclosure without antenna at the point mm x mm x 60 mm As can be seen from Fig. This indicates that the patterns and the number of apertures only affect the level of attenuation to which EM field propagating through apertures is exposed. As expected, the level of SE decreases with the increase of area covered by apertures.

However, as it will be shown next, these results deviate to some extent from the case when the receiving antenna is included in the model. For calculating SE when the receiving dipole antenna is included in the numerical model of the enclosure by using the compact wire model described in section II, the antenna is modelled as z-directed 80 mm long wire having the diameter of 1. Its position within the enclosure is defined by points mm, mm, 0 mm and mm, mm, mm.

The numerical results for the SE of enclosure, shown in Fig. In comparison with the case when enclosure is empty, it can be seen that the antenna presence significantly decreases the SE of enclosure in the whole frequency range. Results for SE of enclosure, with one aperture of dimension 50 mm x 10 mm, at the point mm x mm x 60 mm Fig. Results for SE of enclosure, with one aperture of dimension 50 mm x 30 mm, at the point mm x mm x 60 mm Fig. Numerical results for the SE of enclosure with antenna at the point mm x mm x 60 mm Fig.

Results for SE of enclosure, with two apertures of dimension 50 mm x 10 mm, at the point mm x mm x 60 mm It can be seen that the presence of the receiving antenna significantly reduces the shielding efficiency of the enclosure, as the SE level is always lower in comparison with the case when the enclosure is empty. Although the presence of antenna is neglected in the circuital approach, the SE values of enclosures with one and two 50 mm x 10 mm apertures are lower than the numerical results obtained.

For the enclosure with one 50 mm x 30 mm aperture dimensions, the results obtained by circuital approach practically coincide with the numerical results obtained for an empty enclosure. Also, there is a tendency to slightly shift the resonant frequencies of enclosure. Finally, the rectangular enclosure with apertures on multiple sides for oblique incident plane wave is considered. Numerical results for SE at the point mm x mm x 60 mm obtained by using TLM method with and without compact wire model and results obtained using circuital approach at the same point are shown in Fig.

Two groups of two rectangular apertures of dimension 50 mm x 10 mm are placed on the adjacent enclosure walls while an obliquely incident wave with the azimuth angle 60, elevation angle 90, and polarization angle 30 is used as excitation. It can be seen that the presence of the receiving antenna significantly reduces the SE of the enclosure and that results obtained using circuital approach are lower than the numerical results. Results for SE of enclosure, with two groups of two rectangular apertures of dimension 50 mm x 10 mm placed on the adjacent enclosure walls, at the point mm x mm x 60 mm IV.

The given example confirm that the antenna presence affects the EM field distribution inside the enclosure and thus affects the SE level results, as well as the location of resonant frequencies. The results of circuital approach do not coincide in all cases with the numerical simulation results, but this fast analytical method can be used for approximate calculation of SE of enclosures.

Future research will comprise numerical estimation of antenna impact on SE for given antenna dimensions and also modification of the circuital model in order to include the presence of the receiving antenna, since it is a real element in the measurement process. Park and H. Eom, Electromagnetic penetration into a rectangular cavity with multiple rectangular apertures in a conducting plane IEEE Trans. EMC-0, no. Robinson, T.

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Benson, C. Christopoulos, J. Dawson, M. Ganley, A. Marvin, S. Porter, D. Shim, D. Kam, J. Kwon, J. Kunz, R. Nie, P. Du, Y. Yu, Z. Scientific- Professional Symp. B-I-6, pp. Milutinovic, T. Cvetkovic, N. Doncov, B. Milovanovic, Analysis of enclosure shielding properties dependence on aperture spacing and excitation parameters, in Proc. IEEE Conf. Wlodarczyk, V. Trenkic, R. Scaramuzza, C. The aim of this paper is to analyze transmission parameters such as propagation loss, diffraction fading, attenuation due to atmospheric conditions etc. Reliability of radio-relay links, depending on bandwidth channel, digital modulation types and bit rate, is considered using the simulation programme Pathloss.

Annual rain and multipath unavailability are analyzed at the frequency of 18 GHz band on two links and four bandwidth channels. These results are particularly important in the design of transmission links over Wireless Internet. Keywords Radio-relay, reliability, rain unavailability, multipath I. These methods have been developed for the purpose of predicting the most important propagation parameters for radio-relay links [1].

In design of line-of-sights of radio-relay links several propagation parameters must be taken into consideration. These parameters are: diffraction fading due to an obstacle on terrain path, attenuation in the atmosphere, fading due to atmospheric multipath, fading due to multipath arising from the surface reflection, attenuation due to precipitation, variation of the angle-of-launch at the transmitter terminal and the angle-of-arrival at receiver terminal due to refraction, reduction in cross-polarization discrimination in multipath or precipitation conditions and signal distortion due to frequency selective fading and delay during multipath propagation.

The probability of appearance of these events is of great importance in consideration of reliability of line-of-sight radio-relay systems. All these parameters are calculated in designing of terrestrial radio-relay links to points which are used to provide wireless Internet. From these points wireless Internet is distributed at standardized frequencies to end users. Outage is the loss of ability of a device to perform a required function []. The reliability of a radio-relay system depends on equipment and connection availability. An estimation of the equipment availability is used in order to statistically predict when the equipment or radio-relay system will be unavailable due to unintentional malfunctions on radio-relay devices.

Let us consider the influence of connection unavailability on the reliability of the entire system. The availability objectives, availability ratio AR and mean time between outage Mo , and reciprocal outage intensity OI needed for design purposes are given in ref. The availability objectives applicable to fixed wireless link of length Llink, can be derived from the values given in [3]. The variables in equations 3 and 4 are given in ref.

These values correspond to AR of The length is in the range of km. The results of the analysis are displayed in Figs. Unavailability, 7 MHz bandwidth Fig.. Unavailability, 56 MHz bandwidth number of events per year OI 10 and the mean time between unavailability events Mo min. Selective fading is calculated based on equipment signature for a given radio-relay equipment made by Ceragon [6].

The signature parameter definitions and specifications of how to obtain the signature are given in recommendation ITU- R F. It can be concluded from 1 and that the selective fading is more pronounced at wider channel 56 MHz and higher bit rate, and that by reducing the transmitter power the multipath unavailability increases drastically, as is the case with rain unavailability.

Unavailability, 7 MHz bandwidth Fig. Unavailability, 14 MHz bandwidth Fig. Unavailability, 56 MHz bandwidth QPSK modulation has a lesser multipath and rain unavailability for the same bit rate, even though it uses a wider channel. This fact has to be taken into proper consideration when radio-relay connections of high reliability are designed. Unavailability vs. Hop length was The results from Table I are displayed in Figs. These dependencies must be taken into consideration in order to enable reliable operation when links capacity is designed, since it is obvious that radiorelay availability strongly affects the operation routine.

This is particularly the case with QAM modulation at high bit rates. The calculations have been performed at 19 GHz frequency with vertical polarization. Multipath and rain unavailability dependence on bandwidth, modulation type and The reliability of radio-relay system due to the unavailability of link connection as a function of emitting antenna TX power, bit rate, bandwidth and modulation type has been analyzed, in purpose of study data transmission. Two link connections in the range of 18 GHz with vertical polarization have been discussed. The calculations were performed based on the algorithm defined in ITU-R recommendations with the use of a commercial software packet, while the characteristics of radio-relay equipment were given by manufacturer Ceragon for each bandwidth and modulation.

During calculations of selective fading equipment signature has been used in order to determine precisely the -way multipath unavailability. The dependence of unavailability on the emitter TX power has been analyzed and it was shown that the unavailability drastically increases when TX power is reduced.

Based on tables and figures displayed in the paper a large dependence of rain and multipath unavailability on the bandwidth can be noticed. The unavailability is significantly depended on the modulation type. The unavailability at QPSK modulation for the same bit rate and different bandwidths is considerably lesser than is the case with other types of modulation.

In conclusion, based on presented tables and figures, optimal equipment, which satisfies international standards, for a specific radio-relay internet connection can be designed. This can be useful for internet providing to the distance places. Figures 4a and 4b show plots of unavailability dependence on modulation type for different bandwidths. It can be seen that by increasing bandwidths multipath and rain unavailability also increase.

Multipath unavailability increase is significant at 56QAM modulation, i. As a consequence of this problem, special care must be taken when designing link connections of large capacities. The analysis of unavailability of both hops considered in this paper has shown the same dependence on bandwidth, modulation type and bit rate. ITU-R P.

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They are based on the application of technologies DWDM, frequency stacking, PON, FTTH in order to increase the bandwidth efficiency of both the downstream and upstream paths, to reduce the existing asymmetry between them and to improve the quality of services. Criteria to choose the appropriate components such as laser transmitters, DWDM multiplexers, optic amplifiers etc.

The signals are conveyed from the hubs to the optical nodes over the optical fibers. In the optical nodes the optical signals are transformed into electrical ones. After that the signals are distributed to the subscribers by a coaxial distribution system. Hence, previously built coaxial distribution systems were combined through the usage of optical rings and in this way the subscriber service was localized in one headend.

Cable distribution networks are bi-directional: that makes it possible for additional services such as Internet access, VoD, VoIP etc. Signals transmission over the cable network of a CATV system worsens the quality of service QoS due to noise and distortions inherent to the active devices in the HFC system such as laser transmitters, optical receivers, optical and RF amplifiers. The level of both noise and unwanted spurious signals depends on the parameters of the HFC network components, the dynamic range of RF signals, number of channels, optical modulation depth etc.

Ohridski Blvd, Sofia , Bulgaria, E-mail: jordanova tu-sofia. Ohridski Blvd, Sofia , Bulgaria, E-mail: dobrev tu-sofia. Ohridski Blvd, Sofia , Bulgaria, E-mail: kld tu-sofia. Hence, designers should aim at shortening the coaxial part of HFC networks. The extension of the territorial range of CATV systems and the services package available, and the increasing of the number of their users demands new architectural solutions for building the cable distribution network.

The aim is to increase the system s throughput and to improve the quality of the provided multimedia information. Today, DWDM systems are being deployed to provide network segmentation and increased bandwidth. Additionally multiplexing in the RF domain is also being used in the upstream passband to increase bandwidth efficiency []. The best solution for reducing noise and signal distortion in the amplifiers used to compensate the attenuation in the cable is to move towards a fully passive cable distribution networks such as fiber to the curb FTTC and fiber to the home FTTH [].

This paper presents architectures of CATV systems, based on the combination of these technologies. Two frequency bands are provided for signal transmission from the headend to the subscribers: 11 MHz to MHz for analog video broadcasting and MHz to 86 MHz for narrow casting services data, voice and digital video. The RF signals are transferred over the optic fiber by means of optic carriers whose wavelength may be nm or nm while with DWDM the wavelengths can be chosen from the wave range recommended by ITU from to nm.

The step of the ITU grid is 0. This is done to avoid the appearance of nonlinear distortions due to fourwave mixing FWM. A specific feature of these architectures is that they implement DWDM technology to transmit interactive downstream channels from headend to hub over one optical fiber. The analog transmitter and the ITU transmitters in the headend can be regarded as externally modulated sources that comprise a distributed-feedback DFB laser coupled to a Mach-Zehnder modulator. Outputs from the ITU transmitters are multiplexed and transported to the hub.

To compensate for the loss within the optical channel erbium-doped fiber amplifier EDFA is used. Forward path configuration 1 Node The first DWDM architecture has been developed for the nm range where signals coming over the commonaccess and the interactive channels are combined at the hub in the optic range. The analog transmitter output is optically amplified to a saturated level of about 17 dbm and transmitted through a shared fiber to the hub, amplified again and split into a number of outputs that matches the number of targetedservices wavelengths.

After splitting, the analog signal is combined with the QAM wavelengths and that combination is again split to serve the number of optical nodes for which the given wavelength is targeted. There may be multiple nodes targeted per wavelength, especially in the early deployment stages when subscriber take rates are low corresponding to a low bandwidth requirement per node. At the hub, the signals are returned to RF by means of optical receivers.

Combined at RF, both broadcast and narrowcast signals drive the nm lasers. Essential to the shared use of fibers is a means by which to combine incoming signals at the transmit end and to separate them at the receiving end. The trade-off is that a broadband optical combiner e. Thus, the decision as to whether to use wavelengthspecific or broadband combining at the transmit end of a WDM link must be driven by the consideration of the overall link design.

At the receiving end, however, there is no alternative to using wavelengthspecific demultiplexers if the signals are to be detected separately. Peer-to-peer applications consume a lot of upstream bandwidth. Furthermore, the large picture, audio, and video files being uploaded today demand higher speeds. This pressure on upstream bandwidth has generated several technologies for splitting the node in the upstream direction, without using more fibers to get data back to the hub or headend.

Splitting the node also helps with noise-funneling issues. The return channel capacity may be increased by using a block conversion system, which takes several return paths and converts all except one to a unique block of frequencies. The blocks are combined and transmitted to the headend using one return optical transmitter. Single conversion is used for economy. For other applications, double conversion might be used if it is necessary to put many blocks close together, but the cost is higher.

Figure 3 illustrates block conversion in a common configuration that uses a MHz return optical transmitter. Up to four return paths may be accommodated. The first is coupled directly to the optical transmitter. The other three are up-converted to other frequencies up to about 40 MHz and combined with the unconverted spectrum to supply signals to a return transmitter. At the headend, the process is reversed, developing four individual 5 to MHz spectra to be supplied individually to receivers.

Alternatively, it is possible to build return receivers that tune to 40 MHz, eliminating the need for down-conversion in the headend. Block conversion at an optical node An alternative to block conversion is dense wave division multiplexing DWDM. Each branch coming back to the node is supplied to a different optical transmitter operating on a different wavelength. A DWDM is used to combine the wavelengths on a single fiber for transmission to the headend or hub. A second DWDM is used at the headend or hub to separate the individual wavelengths before they are supplied to individual optical receivers.

There are two configurations currently being investigated to combine these two technologies. The difference is the location of the ITU grid transmitters. In the first configuration these transmitters are located at the hub. In the second configuration, illustrated by Fig. Fed by the RF spectrum from the upconverter the individual wavelengths are transmitted back to the hub. A third technology for combining reverse path signals has also been developed: digitize the return band at the node and transport the digitized signals to the headend. At the headend, the digitized signals are converted again to RF in order to allow interface with legacy headend systems.

RF inputs Most involve a passive optical network PON , which runs one feeder fiber from the central office to a passive terminal, then distributes the transmitted signals over distribution fibers to each of typically 16 to 3 optical nodes. Fiber-deep architectures, such as FTTC and FTTH, are driven by the improved quality of signals delivered over fiber-optic plant as compared to those delivered over coaxial plant, by improved reliability due to fewer devices in the network, and by improved bandwidth.

Because the multiple services analog and digital video, cable modem-based Internet access are multiplexed using separate RF subcarriers, the delivered signals are compatible with existing consumer appliances, and demultiplexing the desired information from the full data stream is very simple. Furthermore, this RF subcarrier multiplexing gives the network operator the ability to gracefully evolve the service mix over time if, for example, the operator wishes to replace some or all of the analog channels with digital services.

In such a system, the outputs of the two diplexer low-pass sections are individually digitized in the two analog-to-digital converters ADC , and then applied to a TDM multiplexer, which alternately passes data from one ADC and then from the other. The data are serialized and supplied to a digital transmitter for transport to the headend.

At the headend, the data is demultiplexed into two signal streams, which are converted to RF in digital-to-analog converters DAC. Fiber-deep architecture In Fig. The local headend typically receives video signals in baseband format from a central primary headend, which receives them from a satellite or from local broadcasters. At the local headend, these signals are converted to AM-VSB format for analog video and to QAM format for broadcast digital video and transported to hubs, each serving roughly 0 subscribers, via a pathredundant supertrunking ring that is, each output of the local headend optical splitter connects to a different hub on the ring via a dedicated fiber, and two connections per hub are made one clockwise around the ring and one counterclockwise for redundancy.

IP data and narrowcast video channels are typically carried together from local headend to hub over separate fibers using synchronous optical network SONET. Targeted services TS are passively combined with broadcast services at the hub and then the downstream signal DWDM-based fiber-deep architecture is amplified and split multiple times. Inserting the TS channels at the hub reduces the sharing of the available TS bandwidth.

Moving the TS insertion point toward the output of the hub reduces bandwidth sharing. Unlike HFC, which uses dedicated downstream and upstream feeder fibers to connect the hub to remote optical nodes, this architecture employs PONs that branch out from each hub and terminate at ONs. Each PON carries bidirectional signals via 1. Coax drops to subscriber homes connect directly to existing in-house coax so that existing customer premises equipment cable-ready TVs, STBs, cable modems, and IP telephones can be connected to the network.

In order to increase the system s capacity DWDM can be used to deliver targeted services.

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DWDM allows to move the TS interfaces CMTSs, video modulators, video servers, telephony bandwidth managers, and advertisement-insertion equipment back to the headend, thereby centralizing the operation and maintenance of the network and saving valuable floor space in hubs. Typical for this architecture is that the TS transmitters are located in the local headend and each is fixed at a controlled wavelength. These sources are multiplexed onto one or more fibers and transported to the hub.

At the hub, the wavelengths are separated and each one is inserted onto a different branch of the network. Thus, by putting a channel on a specific wavelength or a set of wavelengths, it can be targeted to a specific segment of the network. In this case the TS insertion is done optically with a x combiner in the hub. These independent signals, or targeted services, include internet data streams, telephony, requests for and delivery of video-on-demand, near-video-on-demand, and analog channels reserved by franchise requirements for public, education, and government use.

The presented fiber-deep CATV architectures are superior to HFC in that they offer more total bandwidth, both upstream and downstream, which can be shared among fewer homes. By limiting the coax plant to a small passive run in FTTC, or eliminating it altogether in FTTH, minimizing of noises and nonlinear distortions is achieved, which leads to improvement of QoS.

Optical signals on separate wavelengths interact as they travel through the fiber, and those interactions generate various levels of crosstalk. Depending on the parameters of a given link, these mechanisms can have a serious effect on recovered RF signal quality. As our analysis shows, the quality of discrete components in general, and of the wavelength demultiplexer in particular, is typically the limiting factor in achieving acceptable low levels of crosstalk interference.

Large, J. Farmer, Broadband access networks, Elsevier, Inc. Farmer, D. Large, M. Measurements are carried out in an anechoic chamber employing a horn antenna at the transmitting site and a rectangular 4 4 microstrip patch antenna array at the receiver. Measurement process is briefly described as well as the equipment and software used for that purpose.

Frequency dependency of DOA estimates is analysed. Phase differences between signals collected by the array elements make it possible to calculate DOAs, and phase differences between the consecutive frequencies of the complex envelope at each element allow estimating the delays [1]. Using DOA estimation algorithms, both quality of received signals and capacity of mobile communication systems can be significantly improved.

Further, there are a number of location-estimation techniques that are also based on DOA estimation. Employing two or more base stations, and adaptive arrays equipped with DOA estimation algorithms, they are able to determine the transmitter location by calculating the intersection of estimated DOAs from different base stations.

This algorithm is well-known of its super-resolution capability as it provides highly-accurate DOA estimates. Being a subspace-based algorithm, it performs eigen-decomposition of spatial covariance matrix that is formed from the received signals at different elements of an antenna array. In this paper, a system and procedures to provide these data experimentally are described as well as the results of D MUSIC algorithm. To provide hardware simplifications of a data acquisition system and to avoid requirements for a large number of receivers one for each array element , frequency-domain channel sounding technique is employed.

Measurements are based on the VNA Vector Network Analyzer that provides frequency response of the radio channel established between a transmitting and a receiving antenna [3]-[5]. There is only one receiver that is shared in time by use of RF switch matrix. In this way, signals to form a spatial covariance matrix are provided, and the matrix is further processed by D MUSIC algorithm.

Measurements are done for three positions of the transmitting antenna in elevation, for a number of azimuth angles and frequency points. In our future research work, measurement data verified by MUSIC algorithm here will be used to further increase an efficiency of neural network models developed for fast and accurate DOA estimation [6].

System setup for measurements as well as the antennas and calibration procedure performed in the anechoic chamber are described in Section III. Measurement scenario and analysis of measurement results are given in Section IV. Section V contains conclusions remarks. Array element in the origin of the spherical coordinate system is taken as the reference one. Phase differences of signals received at all other array elements are defined relative to the reference. Further, it is supposed that the URA is placed in yz-plane, with distance between adjacent elements in y- and z-direction of dy and dz, respectively.

On the basis of Eq. To determine the angular positions of K sources, a particular property of the sub-spaces spanned by the eigenvectors related to the large and the small eigenvalues of the R matrix is exploited. The beam-steering vectors virtually pointing towards the sources are linear combinations of the eigenvectors related to the large eigenvalues. Hence they must be orthogonal to the noise subspace spanned by the eigenvectors related to the small eigenvalues in the R matrix.

In case the orthogonality condition is fulfilled, the Eq. Therefore, short distance measurement of radio channel characteristics, based on the use of VNA, represents very attractive measurement technique because of very simple implementation requirements, relative flexible system, and ability to track system error [1]. By the use of frequency sweep technique wide dynamic range can be provided for measurements.

Further, it is possible to directly measure absolute losses due the path loss between two observed antennas. On the other side, disadvantage of this measurement method is slow measurement time and requirements for long cables to transmit the referent signal what limits its use on short-distance links with relatively stationary channel.

The measurement system shown in Fig. Frequencychannel sounding is carried out by sweeping a set of narrowband sinusoid signals through a frequency band. The VNA operates in the transfer function mode where one of its ports serves as the transmitting port and the other as the receiving port. Scattering S 1 - parameter is used to express the complex frequency channel transfer function.

VNA sends a frequency tone f through the channel and channel transfer function is represented as S 1 f. At the receiving site, each array element is through the coax cable connected to the appropriate input of RF switch matrix Fig. The signal from the matrix is then guided to the second port of the VNA. Array elements are switched sequentially, where only one array element is active in a moment while other elements are present as dummy elements. Measurements at the network analyzer are averaged 16 times. At each angular position in the anechoic chamber 15 snapshots are recorded.

Existing software for standard antenna measurements, SPAM 3D, is upgraded to implement RF switch matrix as a new instrument in the measurement setup. Positioner in the anechoic chamber is able to rotate in azimuth only, in the 1 16 To make measurements for different elevation positions it is necessary to change the height of the transmitting antenna.

Antennas As a transmitting antenna, a double-polarized quad-ridged horn antenna model no. A , operating in the frequency band. The gain of the antenna is around 7 dbi at. At the receiving site, a rectangular antenna array is positioned, composed of 16 4 x 4 microstrip patch antennas with resonant frequency at. Measured return loss of a single patch is db, and measured gain is 6.

Transmitting antenna mounted on the tripod a , receiving array with RF switch matrix behind placed on the antenna tower b B. Calibration of a system Calibration procedure is necessary to remove errors caused by the source, and more important, to eliminate all frequency dependent effects of a measurement system such as reflections in cables and connectors. The accuracy of the calibration procedure efficiently sets the dynamic range of the measurement system.

To measure a transmission, as it is case with the characterization of a radio channel, response calibration is needed. Calibration results are then used to correct system errors of frequency response [1], [3]. Transfer function S 1 of the complete system is measured in the anechoic chamber using reference antennas on mutual distance of 5.

Calibration data are recorded for each channel of RF switch matrix separately, and used later in the post-processing to normalize measured data. It is known that any change between the calibration and test setups can cause errors in measurement results such as multiple reflections on transmitting and receiving site due to the impedance change when antenna elements are switched and changes of cable characteristics due to flexures.

To minimize such errors it is very important not to move any instrument or cable while the measurement is running. Receiving antenna array is rotated to a position in azimuth. This procedure is repeated for all 16 channels. This is done for all required azimuth positions.

Reference DOAs are 6, 1. Solid lines are plotted for DOA estimates and vertical dotted lines present the expected angular positions. It can be concluded that measured values are in good agreement with reference DOAs. On the other side, there is an error of 1. The largest error is 4.

All measurements are carried out in an anechoic chamber taking into the consideration only the direct signal between the transmitting and the receiving antenna. As it is demonstrated, a element microstrip patch array has good performance for D DOA estimation. To provide more accurate D DOA estimates, mutual coupling between array elements should be compensated performing careful calibration procedure for the array.

Angular positions in azimuth solid lines - DOA estimates, dotted vertical lines actual positions. Street, L. Lukarna and D. Antennas Propag. Macedo, M. Dias, R. Vieira, J. MacQdo and G. Siqueira, Mobile Indoor Wide-Band 1. Vehicular Technology Conference, vol. Hajian, C. Coman and L. Hafiizh, S. Obote and K. As a measure of quality used optical Q factor. On the basis of obtained results is discussed how to modify the transmission quality and behavior of the thermal noise with a variation of numbers amplifying section and the quantity of flow per channel. Graphically is shown change Q factor for different values of the thermal noise, the number amplifying section and length amplifying section.

At least attenuation in the optical fiber is achieved by applying the wavelength of nm, i. WDM systems allow expansion of existing capacity without laying additional fiber optic cables. The capacity of the existing system is expanding using multiplexers and demultiplexers at each end of the system [3], [4]. To enable 16 channels on one fiber, CWDM uses the entire frequency range between the second and third optical window nm and nm.

For successful transmission of optical signals over long distances is using the EDFA amplifiers. Weak signal enters in the erbium doped fiber in which light is injected using the laser to pump. This light excites erbium atoms to release stored energy as additional light of wavelengths around nm.

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